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ISL85033
Regarding transient response needs, a good starting point is to
If capacitors other than MLCC are used, attention must be paid to
I RMS
------------ = D – D 2
determine the allowable overshoot in V OUT if the load is suddenly
removed. In this case, energy stored in the inductor will be
transferred to C OUT causing its voltage to rise. After calculating
capacitance required for both ripple and transient needs, choose
the larger of the calculated values. The following equation
ripple and surge current ratings.
Io
where D = V O /V IN
(EQ. 10)
determines the required output capacitor value in order to
achieve a desired overshoot relative to the regulated voltage.
The input ripple current is graphically represented in Figure 45.
V OUT 2 * ( V OUTMAX ? V OUT ) 2 – 1 )
I OUT 2 * L
C OUT = -------------------------------------------------------------------------------------
(EQ. 8)
0.6
0.5
where V OUTMAX /V OUT is the relative maximum overshoot
allowed during the removal of the load. For an overshoot of 5%,
0.4
V OUT 2 * ( 1.05 2 – 1 )
the equation becomes:
I OUT 2 * L
C OUT = ----------------------------------------------------
(EQ. 9)
0.3
0.2
The graph in Figure 44 shows the relationship of C OUT and %
overshoot at 3 different output voltages. L is assumed to be 7μH
0.1
and I OUT is 3A.
0
0
0.2
0.4
0.6
0.8
D
FIGURE 45. I RMS /I O vs DUTY CYCLE
80
A minimum of 10μF ceramic capacitance is required on each VIN
60
40
20
12V OUT
5V OUT
3.3V OUT
pin. The capacitors must be as close to the IC as physically
possible. Additional capacitance may be used.
Loop Compensation Design
ISL85033 uses a constant frequency current mode control
architecture to achieve simplified loop compensation and fast
loop transient response.
0
1.02
1.04
1.06
1.08
1.10
The compensator schematic is shown in Figure 47. As mentioned
in the COUT selection, ISL85033 allows the usage of low ESR
V OUTMAX /V OUT
FIGURE 44. C OUT vs OVERSHOOT V OUTMAX /V OUT
Current Sharing Configuration
In current sharing configuration, FB1 is connected to FB2, EN1 to
EN2, COMP1 to COMP2 and VOUT1 to VOUT2 as shown in
Figure 3. As a result, the equivalent gm doubles its single
channel value. Since the two channels are out-of-phase, the
output capacitor. Choice of the loop bandwidth f c is somewhat
arbitrary but should not exceed 1/4 of the switching frequency.
As a starting point, the lower of 100kHz or 1/6 of the switching
frequency is reasonable. The following equations determine
initial component values for the compensation, allowing the
designer to make the selection with minimal effort. Further detail
is provided in “Theory of Compensation” on page 20 to allow fine
tuning of the compensator.
Compensation resistor R1 is given by Equation 11:
g m V FB
frequency will be 2X the channel switching frequency. Ripple
current cancellation will reduce the ripple current seen by the
output capacitors and thus lower the ripple voltage. This results
in the ability to use less capacitance than would be required by a
single phase design of similar rating. Ripple current cancellation
also reduces the ripple current seen at the input capacitors.
2 π f c V o C o R T
R 1 = ---------------------------------
which when applied to ISL85033 becomes:
R 1 [ k Ω ] = 0.008247 ? f c ? V o ? C o
(EQ. 11)
(EQ. 12)
Input Capacitor Selection
To reduce the resulting input voltage ripple and to minimize EMI by
where C o is the output capacitor value [μF], f c = loop bandwidth
[kHz] and V o is the output voltage [V].
C o × V o × ( 10 )
C o × R c × ( 10 )
I o × R 1
C 1 = ---------------------------------------- , C 2 = ----------------------------------------
forcing the very high frequency switching current into a tight local loop,
an input capacitor is required. The input capacitor must have adequate
ripple current rating which can be approximated by the Equation 10.
Compensation capacitors C1 [nF], C2 [pF] are given by
Equation 13:
3 6
R 1
(EQ. 13)
19
FN6676.6
February 23, 2012
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